8.2.1   C-Shaped Antenna with Switchable Wideband Frequency Notch

The number of bands are increasing day by day with multiple high-speed services. These services need a high bandwidth and need to be separated from each other to avoid interference between channels. There are two primary design requirements for this. The first is an antenna design with frequency notch and the second is an antenna design with high cross-polarization attenuation. The notched frequency band refers to frequency response with a small stopband and cross-polarization refers to the orthonormal orientation of electric fields (vertical and horizontal). The adjacent bands in a satellite are isolated by notched frequency bands and polarization, that is, if the first channel is vertically polarized, the second will be horizontally polarized and the third will again be vertically polarized, and so on. This provides isolation among various channels. The antennas designed for multibands with a stopband in between are known as frequency notched antennas. For instance, three bands are allocated for Wireless Local Area Network (WLAN) 2.4 GHz (2,400–2,484 MHz), 5.2 GHz (5,150–5,350 MHz), and 5.8 GHz (5,725–5,825 MHz). Worldwide, the WiMAX system operates at the frequency bands of 2.5 GHz (2,500–2,690 MHz), 3.5 GHz (3,400–3,690 MHz), and 5.8 GHz (5,250–5,825 MHz). Another high-speed WLAN service of 1 Gbps speed is under development (IEEE 802.11.ac), which is to operate around 5 GHz.

Many techniques have been proposed for designing band-notched antennas. The simplest method is to etch slots on the radiating patch of the antenna, such as U-shaped slots [1], L-shaped slots [2,3], and V-shaped slots [4]. An effective way of achieving a UWB compact frequency notched antenna is the use of a split ring resonator (SRR) [5, 6 and 7]. A square split ring resonator is used in [5], while in [6], a slot-type SRR has been used to design an ultrawideband notched antenna. Multiple notched frequency bands have been implemented with an SRR below feed line in [7], with simple slots in [8], and using a half-mode substrate in [9]. A frequency notched printed slot antenna has been implemented in [10], while a compact antenna is designed using fractal geometry in [11]. Instead of using the conventional fractal geometry such as Koch curves, Sierpinski triangles, and Minkowski fractals, a Koch-curve-shaped slot has been employed to achieve a compact geometry with frequency notch characteristics.

The coaxial probe feed is one of the most popular feeding techniques for electrically thick substrates, but the inductance of the probe creates impedance mismatch and causes low bandwidth. The inductance of the probe is compensated for by cutting slots on the patch, using an L-shaped strip feed [12] or by introducing a capacitive feed strip [13,14]. But these configurations create theoretical complexity in terms of analysis apart from the mechanical alignment issues while assembling and hence may increase the production cost. Capacitive feed provides a simple and ultrawide impedance matching by compensating for probe inductance with its capacitance.

A C-shaped antenna may be described as an RMSA with rectangular notch at one of the edges [15]. This is a very popular technique to achieve dual band [16] and wideband [17] characteristics.

In this design, a new technique has been presented to achieve a frequency notched antenna. A simpler capacitive coupling technique has been employed to get ultrawideband characteristics. A PIN diode has been used so that the antenna operates in ultrawide band as well as dual band with a frequency notched antenna. The gain and other radiation characteristics are normal, while the cross-polarization attenuation is very high.

8.2.2   Multiband Multipolarized Reconfigurable Circularly Polarized Monopole Antenna with a Simple Biasing Network

In recent years, circularly polarized monopole antennas have attracted a great deal of attention for the current wireless communication system, as circular polarization (CP) plays a very important role in improving the quality of the received signal [18]. These applications include the C-band communication satellite from 5.925 to 6.425 GHz for their uplink and amateur satellite operations in the frequency range of 5.830 to 5.850 GHz for downlinks. The X-band uplink frequency band uses 7.9–8.4 GHz for military communication systems. The traffic light crossing detector operates at 10.4 GHz. In Ireland, Saudi Arabia, and Canada, terrestrial communication uses the bandwidth of 10.15–10.7 GHz. The usage of multiple antennas for achieving different CP radiations will make the system complex. Hence, as per the demand of the next-generation communication systems, single multiband circularly polarized antennas can be used to reduce the complexity of the system. Therefore, designing such antennas have attracted a great deal of attention among the researchers. However, designing such antennas is challenging when the number of operating CP frequency bands increases. Generally, a monopole antenna generates linearly polarized radiation. Hence, it is difficult to radiate CP waves which were produced by two near-degenerated orthogonal resonant modes of equal amplitude with opposite phase differences. In [19], CP is generated in the triangular patch by using a Koch curve. The axial ratio bandwidth is about 1.3%. A triple proximity-fed microstrip antenna gives the CP with an axial ratio bandwidth of 0.70% for the L-band [20]. In [21], an elliptical microstrip antenna with proximity coupling is used to excite CP waves with an axial ratio bandwidth of about 0.85%. In [22], a simple circularly polarized antenna is presented. The CP operation is obtained by using an open slot having an open width at the lower side of the model. In [23], a slot antenna with straight feed is presented. An SRR-inspired structure is used to obtain the circular polarization. Wideband high gain circularly polarized antennas are proposed in [24]. The use of feed and parasitic patches makes the structure complex. Many CP monopole antennas were proposed for various applications in wireless communication [25, 26, 27 and 28]. In [25], an asymmetrically shaped radiator fed by a microstrip line and a limited ground plane is presented. In [26], the CP is achieved by loading four cylinders that are perpendicular to the substrate of the microstrip antenna near the edge of the circular patch. In [27], by using a rectangular dual-loop technology and tuning the separation between the ground plane and antenna, the monopole antenna produces a CP. In [28], an asymmetric antenna geometry is used to obtain the wideband CP. However, CP monopole antennas mentioned above focus on single CP band operation. Dual band CP has been investigated in [29, 30, 31, 32 and 33]. The antenna presented in [29] composed of a partial ground plane and a Y-shaped radiating patch that consists of two unequal monopole arms and a modified circle. The dual CP is obtained with two unequal monopole arms and a modified circle with an axial ratio bandwidth of 3.8% and 6.8%. In [30], dual CP is obtained by embedding an inverted-L slit in the ground plane. A halved falcate-shaped dual-broadband CP printed monopole antenna is proposed in [31]. To generate dual CP, two halved falcate-shaped antennas were used to generate orthogonal modes and three stubs in the ground plane were used to give 90 degree phase difference. A novel monopole antenna with dual CP consisting of a radiating patch composed of an annular-ring linked by a square ring over the corner and a ground plane with embedded rectangular slit was proposed in [32]. In [33], the dual band CP operations are realized by using two parallel monopoles – one curved monopole and one fork-shaped monopole – and a crane-shaped strip is placed on the ground plane. A dual-feed, dual-band-stacked, CP patch antenna system is presented in [34]. The use of dual feed and stacking gives the structure complex. Triple band CP radiations are presented in [35,36]. In [35], a hexagonal slot antenna with L-shaped slits is presented with a narrow 3 dB axial ratio bandwidth of 1.7%, 3.86%, and 5.23%, while 3 dB axial ratio bandwidths of 9.8%, 4.6%, and 2.8% have been achieved in [36]. The complexity of these two designs [35,36] may hinder the integration of antennas in different applications. In [37], an inverted U-shaped radiator rotated by 45° around the horizontal axis is used to generate the triple band CP.

Recently, reconfigurable CP antennas have attracted significant attention. The PIN diodes are switched in different states to obtain the tunable property of the antenna [38]; this technique is also used to generate a reconfigurable CP microstrip antenna [39]. The authors in [40] proposed a dual feed microstrip patch antenna with frequency and polarization reconfigurability. The frequency and polarization reconfigurability are achieved by using six PIN diodes. The design is complex because of the six PIN diodes and complex biasing circuits. A frequency- and polarization-reconfigurable antenna using PIN diodes is designed in [41]. PIN diodes are commonly used as switching devices for RF and microwave application systems, as they have the advantages of low insertion loss, good isolation, and low cost [42]. It has been observed from literature reviews that mainly two considerable problems are often encountered in reconfigurable CP antenna designs. The excessive diodes will lead to a complex dc-bias network [19,43], providing independent bias for each diode by some special mechanism, such as using capacitors [44].

In this design, a very simple monopole antenna with reconfigurable multiband CP operations is presented. The antenna has overcome the abovementioned two problems of the reconfigurable CP antenna. The antenna generates right-hand circular polarization (RHCP), left-hand circular polarization (LHCP), and linear polarization (LP) in different bands with a simple biasing network without making the system complex. By removing the triangular portion at the lower edges of the rectangular radiating patch and embedding the cross-shaped slit at the right lower edge and L-shaped slit in the ground plane, the antenna provides two CP bands with three LP bands in the OFF state and three CP bands with three LP bands in the ON state of the PIN diode. The reconfigurable CP is achieved by using a PIN diode on the L-shaped slit in the ground plane. The multiband CP operation is obtained by controlling the ON/OFF state of the diode. The antenna is suitable for many applications such as traffic light crossing detectors at 10.4 GHz, C-band communication satellites for their uplink from 5.925 to 6.425 GHz, amateur satellite operations for downlinks from 5.830 to 5.850 GHz, terrestrial communication from 10.15 to 10.7 GHz, and X-band uplink frequency band from 7.9 to 8.4 GHz for military communication systems.

8.2.3   Design of C Shape Antenna with Switchable Wideband Frequency Notch

The antenna design is divided into three parts to examine the response at various phases. First, a capacitive-coupled C-shaped antenna is investigated. Two notches are cut symmetrically in the structure to introduce a new resonant frequency in its frequency response, and finally a parasitic rectangular patch is connected to the C-shaped microstrip antenna (CSMSA) via the PIN diode. The ON condition of the diode is implemented through a connection while the OFF condition is without any strip. The final antenna structure is shown in Figure 8.1. Figure 8.2a shows the fabricated antenna when the PIN diode is OFF, while Figure 8.2b shows when the diode is ON. The simulation is carried out on IE3D software and the measurements are taken using an Agilent N5230 vector network analyzer. The antenna is fabricated on an RT Duroid 6002 substrate with a dielectric constant of 2.94 and a loss tangent of 0.0012. The thickness of the substrate is 0.76 mm, which is raised by 6 mm in air. The other dimensions of the antenna are given in Table 8.1.

8.2.4   Multiband Multipolarized Reconfigurable Circularly Polarized Monopole Antenna with a Simple Biasing Network

The proposed reconfigurable multiband circularly polarized microstrip-fed monopole antenna is shown in Figure 8.3. The proposed CP antenna is designed and fabricated on an FR4 substrate of thickness h = 1. 6 mm and relative permittivity ε_r = 4.4. An SMA connector is connected to a microstrip feed line of width W₁ and length L₁, which is connected to an impedance transformer of width W₂ and length L₂. The approximate value of the length of the monopole antenna radiating strip is given by the following formula [30]:

where ε_eff = (ε_r + 1)/2, c is the speed of light, λ₀ is the free space wavelength at the monopole resonant frequency f₀ and ε_eff is the approximated effective dielectric constant. The dimensions of the rectangular radiator of the antenna are L₀ × W₀. Generally, a monopole antenna generates either vertical or horizontal linearly polarized radiations and finds difficulty to generate two orthogonal current components with an equal amplitude and 90° phase differences. CP radiations can be generated by introducing a perturbation segment into a linearly polarized antenna; the linearly polarized orthogonal modes can be converted to LHCP or RHCP. Thus, to meet the desired CP conditions, a triangular portion is cut from the left lower edge of the radiator patch to adjust for the impedance matching and an L-shaped slot is embedded in the ground plane to generate the dual CP radiation [45]. Further, to produce 90° phase differences between the currents at distinct frequencies, a triangular portion is removed from the right lower corner of the monopole radiator and a cross-shaped slit is introduced on the truncated right lower edge of the radiator patch to disturb the surface current on the patch. The antenna exhibits a dual band behavior with two different CP radiations. To design the reconfigurable CP, a PIN diode as the switching component is used. The reconfigurable CP antenna has an L-shaped slot with a PIN diode (SMP1320-079) in the ground plane, as shown in Figure 8.3. When a positive voltage (V₁ = +0.73 V) is applied, a diode acts as a short circuit (ON state) with a small resistance (0.9 Ω). When a zero voltage is applied, the diode acts as an open circuit (OFF state). The geometry of the antenna varied with the ON/OFF state of the PIN diode, providing the proposed antenna with switchable CP bands in accordance with the change in geometry. A dc bias circuit is used to control the ON/OFF state of the PIN diode as shown in Figure 8.3a. A narrow slit in the ground plane is used for dc isolation. These two isolated portions of the divided ground plane are connected to each other in an ac manner by the use of capacitors. Since the bias circuit is located on the ground plane, an RF choke is not required to isolate the bias circuit from the radiation element, resulting in a reduced effect on the radiation by the bias circuit [46]. During the simulation, the ON state of the PIN diode is implemented with a copper link of length 1 mm and width 1 mm. The detailed dimensions of the proposed reconfigurable multiband CP antenna are listed in Table 8.2.

The evolution of various stages involved in the proposed antenna design is shown in Figure 8.4, while the reflection coefficients and axial ratios obtained in each stage are shown in Figures 8.5 and 8.6, respectively. The basic element used in the antenna design is a conventional rectangular patch and ground plane (Ant1). From Figure 8.5, it can be seen that in this case, three resonant modes are obtained at 7.75, 10.42, and 14.92 GHz. Ant1 exhibits the linear polarization (3-dB axial ratio > 3) as depicted in Figure 8.6. Ant2 shows that resonance frequencies shifted downward and operated at multiband frequencies due to modification of the ground plane by embedding an L-shaped slot in the ground plane (Ant2). The dual CP bands at the center frequencies 3.34 and 8.58 GHz are obtained as shown in Figure 8.6 (Ant2). The upper CP band is not within the operating band. Ant3 indicates that the removal of the triangular portion from the lower edges of both sides of the radiator patch provides two wideband operations and three CP operations with a small ARBW as shown in Figures 8.5 and 8.6, respectively. Finally, the radiating patch is modified by cutting a cross-shaped slot on the truncated right lower edge to obtain dual band characteristics and dual band CP behavior with a good ARBW. The proposed antenna is simulated by the Method of Moments–based IE3D simulator. Figure 8.7 shows the fabricated prototype of the antenna.

8.2.5   Characteristics of the C-Shaped Antenna with a Switchable Wideband Frequency Notch

A capacitive coupled C-shaped antenna is designed to operate in the frequency band of 5–8 GHz. The antenna exhibits two resonance frequencies. To achieve a sharper frequency response, two notches are cut symmetrically in the C-shaped microstrip antenna. Figure 8.8 shows the return loss variation of CSMSA with and without symmetrical notches. Clearly, a new resonant frequency is added at the higher side of the response.

After the cutting of notches, a parasitic patch is kept in the etched portion (mouth) of the CSMSA. The effect of putting the rectangular parasitic patch is shown in Figure 8.9. It is clear from the figure that the response remains almost unchanged after placing the parasitic rectangular patch. The slight shift in the response may be due to the mutual coupling between the structures.

After placing a parasitic element to the structure, a PIN diode (model HSMP-3860) is connected between them. The PIN diode is implemented by 1.2 mm× 1 mm through connection. The simulated and measured return loss variation when the PIN diode is OFF is shown in Figure 8.10. There is an acceptable resemblance between them. When the diode is ON, a wide frequency band around 5 GHz is notched, as shown in Figure 8.11. In practical applications, a sharp frequency notch is desirable to isolate the two bands. Here, a simple wide frequency notch is adopted to isolate the two bands. Thus, the antenna can be used both for ultrawide band and for narrow band. In fact, this notch appeared due to a shift in the lower resonant frequency towards the lower frequency spectrum and an impedance mismatch in between. This decrease in frequency was due to the enlarged current path.

The current distribution at old and new (shifted) resonant frequencies is shown in Figure 8.12.

To observe the behavior of the resonant frequency of the newly formed narrow band, the return loss is taken for different lengths of the parasitic rectangle, as shown in Figure 8.13. It is clear from the figure that as the length of the rectangle increases, the resonant frequency decreases, which shows the path length of current lines.

8.2.6   Other Radiation Characteristics

Figure 8.14 shows the gain of the tunable C-shaped microstrip antenna. It is observed from the figure that a better gain is obtained because of the better matching when the diode is OFF. A sharp fall in gain is observed in the stop band when the diode is ON. Figure 8.15 shows the radiation pattern at various resonant frequencies for the ON and OFF conditions of the PIN diode in the E-plane. It is observed from the figure that the radiation pattern is the same for 5.6, 6.5, and 7.7 GHz, which are in the same band of operation. When the band of operation changes (at 4.7 GHz), the main lobe of beam rotates from −10.48° to −30.3°, which is again due to the current path lines flowing in the inner rectangle. Figure 8.16 shows the current lines for 6.5 and 7.7 GHz when the diode is OFF and ON. When the diode is OFF, there is no coupling between the C-shaped antenna and the rectangle. In the ON condition, a little coupling is observed at 6.5 GHz, while being negligible at 7.7 GHz. This is why the shift in the second resonant frequency is the most (Figure 8.11).

8.2.7   Multiband Multipolarized Reconfigurable Circularly Polarized Monopole Antenna with a Simple Biasing Network

The measured and simulated reflection coefficients and the measured and simulated axial ratios with the OFF state of the diode are shown in Figure 8.18a and b, respectively. The proposed antenna exhibits a dual band behavior in the OFF state of the diode having an impedance bandwidth of 10.22% and 83.43% in the frequency range of 3.99–4.42 GHz and 5.84–14.20 GHz, respectively. In the OFF state of the diode, the antenna has successfully achieved two CP bands operation with a 3-dB axial ratio bandwidth of 9.83% and 13.73% in the frequency range of 5.79–6.39 GHz (CP1) and 8.49–9.74 GHz (CP2), respectively, and three LP bands in the frequency range of 3.99–4.42 GHz, 6.40–8.48 GHz and 9.75–14.20 GHz. The measured and simulated reflection coefficients for the ON state of the diode are shown in Figure 8.19a. Figure 8.19b illustrates the measured and simulated axial ratios in the ON state of the diode. In the ON state, the antenna also exhibits a dual band behavior with an impedance bandwidth of 49.10% and 21.31% in the frequency range of 7.20–11.04 GHz and 12.66–15.37 GHz, respectively. When the diode is set to the ON state, a switchable CP antenna is obtained. The antenna attains triple band CP operations with a 3-dB axial ratio bandwidth of 6.49%, 8.45% and 2% in the frequency range of 7.90–8.43 GHz (CP3), 10.08–10.97 GHz (CP4) and 12.87–13.13 GHz (CP5), respectively, and three LP bands in the frequency range of 7.20–7.89 GHz, 8.44–10.07 GHz and 13.34–15.37 GHz. The measured results are within reasonable agreement with simulated results. The differences between the simulated and measured results are due to the fabrication imperfections. They are also the reason that the simulation uses an open gap and a copper strip to represent the equivalent circuits of the PIN diode in the OFF and ON states, respectively. The reflection coefficient is tested by an Agilent5230A vector network analyzer, and radiation performances are measured in an anechoic chamber.

The total gain of the circularly polarized antenna can be given as [28,45,47]

where G_TV is the partial power gain with respect to linearly vertical polarization and G_HV is the partial power gain with respect to linearly horizontal polarization. A linearly polarized standard horn antenna with a calibrated gain is used in the measurement of G_TV and G_HV. The simulated and measured gains with the ON/OFF state of the diode are shown in Figure 8.20. The differences in the measured and simulated results of the gain are due to the loss in the PIN diode and measurement error. The peak gain of the antenna in the 3-dB axial ratio bands are 1.30 and −0.60 dBic at the center frequencies of the CP bands, respectively, for the OFF state, while 1.41, −0.41 and −0.86 dBic at the center frequencies of CP bands, respectively, for the ON state of the diode. The simulated and measured radiation efficiencies of the proposed antenna in the ON and OFF states of the PIN diode are shown in Figure 8.21. The radiation efficiencies remain in the range of 65 ± 5%. The measured and simulated results agree well within the acceptable limit. Figure 8.22a and b plot the simulated and measured normalized RHCP and LHCP radiation patterns at the center frequencies of the CP radiations for the OFF state of the diode, while Figure 8.23a–c plot the radiation patterns at the center frequencies for the ON state of the diode in the two orthogonal planes (E-plane (Φ = 0°) and H-plane (Φ = 90°)). It can be seen that good LHCP and RHCP are observed at the center frequencies of the CP bands CP1 and CP2, respectively, for the OFF state of the diode. In the ON state of the diode, the RHCP, RHCP, and LHCP wave are observed at the center frequencies of the CP bands CP3, CP4, and CP5, respectively.

8.2.8   Radiation Mechanism

In order to understand the operating mechanism and how CP operations are obtained, the simulated current distribution at center frequencies of different CP radiations for t = 0, t = T/4, 2T/4 and 3T/4(T = time period) are shown in Figures 8.24 and 8.25 in the diode OFF and ON state, respectively. At 6.1 GHz, the resultant of the surface current is at 257.5° and clockwise rotation of surface currents at different time frames clearly indicates LHCP. The L-shaped slit in the ground plane and cross-shaped slit in the patch are responsible for generating circularly polarized waves. At 9.1 GHz, the resultant surface current is due to the patch and L-shaped embedded ground plane is about 45° at t = 0 and rotates anticlockwise and the antenna radiates LHCP waves with time. In the diode ON condition as shown in Figure 8.25, at 8.43 GHz, the currents at the L-shaped embedded ground plane are cancelled out by each other, and the resultant current is due to the cross-shaped slit at t = T/4.

The field rotates in the anticlockwise direction, which results in exciting an RHCP radiation. At 10.525 GHz, the cross-shaped slit in the radiator patch is responsible for generating the RHCP radiation, as the surface currents rotate in anticlockwise manner. The clockwise rotation of the resulting surface current is responsible for producing the LHCP mode, at 13.00 GHz. As observed from the above figures, in the ON state of the diode, the surface currents in the L-shaped slotted ground is cancelled out and the resultant current is zero. The only CP is produced due to the asymmetric cross-shaped slit in the patch. Table 8.3 summarizes the sense of polarization of reconfigurable CP bands.

8.2.9   Parametric Study

In order to analyze the operational performance of the presented antenna, the vital parameters are investigated to evaluate their impacts on the antenna. The parameters of cross-shaped slot and L-shaped slit are examined to find their impacts on the antenna performance and are illustrated in Figures 8.26 and 8.27, respectively. The parametric study is carried out by varying the parameters S₂, S₃, S₄, S₅, l₁, and l₂ to examine their effects on the reflection coefficients and axial ratio. It can be seen from Figure 8.26a–e, that by increasing and decreasing S₂, S₃, S₄, S₅, and l₂, little change is observed on the lower band and a very slight change is observed on the upper band of the reflection coefficient. The reflection coefficient is strongly dependent on l₂ as shown in Figure 8.26f. By decreasing the length l₂, the whole band is disturbed and reflection coefficient is above −10 dB.

The changes in CP performances by varying the parameters S₂, S₃, S₄, S₅, l₁, and l₂ are shown in Figure 8.27. The CP performances are slightly affected by the change in the value of S₂, S₄, S₅, and l₂ as shown in Figure 8.27a–d, respectively. The CP bands strongly depend on the value of S₃ as shown in Figure 8.27e. By increasing and decreasing S₃, four CP bands with small ARBW are obtained, out of which two CP bands are not lying in the operating bands. An increase in the length l₁ has little effect on the CP operation at lower and upper bands, while a decrease in l₁shows that the CP bands shifted upward as shown in Figure 8.27f.

8.3.1   Antenna Design

The MED antenna is a combination of an electric dipole and a magnetic dipole. The vertical metallic walls act as the magnetic dipole, which is shorted with the ground plane at one end and the horizontal metallic electric dipole patch in the other end. In the proposed MED antenna, defected semicircular shaped horizontal patches are used as electric dipoles and the folded vertical walls as magnetic dipoles. The detailed geometry of the proposed antenna is shown in Figure 8.28. It is a combination of a minimum rectangular ground of dimension G_L × G_W, a horizontal pair of defected semicircular patches of radius r and two parallel vertically oriented folded walls, which are placed at an optimized distance. The vertical walls are separated by a gap G until height h₂, and the gap of 2 L + G is maintained from height h₁ to h₂. These gaps play a very important role in stabilizing the antenna gain and reduction in the back-lobe radiation. The use of a larger ground plane generally enhances the directivity with restrictions on practical issues, such as installation and bulkiness at the same time. Therefore, a minimum-sized rectangular ground has been used to maintain the compactness of the proposed design; as a result, a rectangular patch of dimension G_L × G_W is formed at height h₂ and a semicircular patch is used at height h₁, which also confirms the maximum height of the structure from the ground plane.

In order to initiate the CP operation, an I-shaped slot is cut in each semicircular patch with the help of a sharp mechanical tool. These slots have a diagonal symmetry. Due to the I-shaped slots and symmetrical folded structure, the proposed design exhibits wideband CP radiation characteristics. The gap between the extended parts of the I-shaped slots from the flat edge of the semicircular patch is 4 mm. The lower and upper parts of the I-shaped slots on the left side as well as the right side of the semicircular structures are extended until the curved edges to widen the 3-dB AR bandwidth.

To excite the antenna, the SMA connector is connected to the Γ-shaped feed line, which is divided into three segments a, b, and c with the uniform width f. It can be seen from Figure 8.28 that two vertical elements and one horizontal element are combined to make this type of feeding structure. The first segment (segment a) is connected to the inner conductor of the SMA connector, which is separated from one of the vertical walls by 1 mm and the other wall by 11 mm air gaps and acts as an air-filled microstrip line. The open end of this line is connected to the horizontal coupling line (segment b) of the feed line, whereas the other vertical segment of the feed line (segment c) is connected to the horizontal coupling line. The outer conductor of the connector is soldered to the ground to complete the feed loop. When the antenna is excited, the horizontal segment of the feed structure couples the energy to the semicircular defected patches and the other vertical segment is used to compensate for the inductive effect of the horizontal segment. By adjusting the size of the feed, proper impedance matching is obtained with a characteristic impedance of 50 Ω. The resonance of the antenna moves upward and downward with variation in the length b and height c. All the dimensions and parameters of the proposed design are shown in Table 8.5.

8.3.2   Parametric Studies

The proposed wideband circularly polarized MED antenna can operate in the frequency range of 3.71–6.91 GHz and exhibits an axial ratio bandwidth of 20.60% in the frequency range 3.71–4.55 GHz. Here, the optimization goal is to achieve the CP behavior as well as wideband operation. The parametric study in terms of its various controlling elements is carried out to obtain the optimum values. During optimization, several important parameters are investigated for optimum performance, such as the radius of the semicircular patch (r), the height of the semi-circular patch (h₁), the height of the metallic patch (h₂), the gap between two magnetic dipoles (G), and the width of I-shaped slots. During the investigation, while one design variable is changed, the other parameters were kept constant. In Figure 8.29, the different values of radius (r) are evaluated to check its impact on the reflection coefficient. It has been observed that the variation in r has a significant effect on the reflection coefficient variation.

The impedance bandwidth gets narrower and lower frequency point is shifted towards the lower side when r is increased beyond 29 mm. It can also be noted here that at r = 31 mm, the antenna exhibits dual band characteristics. Similarly, the bandwidth is reduced, when r is kept less than 29 mm, but the lower frequency point is shifted upward. However, in both circumstances, the upper cutoff frequency either remains same or decreases. Hence, it is confirmed that the optimum impedance bandwidth is achieved at the value of r = 29 mm.

The height of the semicircular patch (h₁) is varied, and the corresponding reflection coefficient is presented in Figure 8.30.

When h₁ is increased beyond 20 mm, the operating band gets distorted, and when h₂ is reduced below 20 mm, the impedance bandwidth gets narrower because of a significant decrease in the second and third resonance frequencies and thereby the upper cutoff frequency. When the height (h₁) is decreased below 20 mm, the whole operating band moves upward but the resultant bandwidth decreases. Hence, the optimum value of h₁ is 20 mm. Here, it is noticeable that lower resonance frequency remains almost unchanged. Figure 8.31 shows the effect of variation in the height of the horizontal rectangular folded patch (h₂), which is varied at the intervals of 0.5 mm. Here, the lower resonance frequency is also unaffected by variation in h₂. When h₂ is increased beyond the design point of 10 mm, the second resonance frequency decreases.

Moreover, the third resonance point is mismatched and hence, the bandwidth decreases drastically. When it is decreased below the design value, the antenna behaves like a dual band antenna because of the mismatch in the operating band. Therefore, it can be projected here that the height of the folded conducting element of the antenna plays an important role in accomplishing the goal of the proposed design and the optimum value is at h₂ = 10 mm.

Figure 8.32 demonstrates the comparison of the reflection coefficient for different values of gap (G) between the two vertical walls.

Maximum impedance bandwidth has been achieved at G = 12 mm as depicted in the figure. A gradual increment in bandwidth is seen up to 12 mm, beyond which the bandwidth gets narrower and band splits into two bands. At G = 13 mm, the bandwidth is reduced, and hence it can be stated that the distance between two walls is also an important factor to obtain a broad bandwidth. The effect of uniform width (S_w) of the I-shaped slot on axial ratio bandwidth is explained in Figure 8.33.

The AR bandwidth has changed significantly for instantaneous values of S_w, while S_ℓ and S_h are kept constant. When S_w ≤ 4.5 mm, the AR bandwidth gets narrower and for S_w = 5.5 mm, the dual CP band response is shown in the figure; out of these, one CP band does not lie in the operating frequency range. The maximum AR bandwidth is achieved when S_w is fixed at 4.5 mm throughout the I-shaped slot. Embedding I-shaped slots in an extended manner with optimized slot dimensions forced the surface currents to rotate in the counterclockwise direction. Hence, it can be concluded that the optimum AR bandwidth is achieved at S_w = 4.5 mm.

8.3.3   Characteristics of the Magnetoelectric Dipole Antenna

The proposed design has been simulated using an Ansys HFSS simulator and the antenna structure is fabricated by using copper sheet only. The thickness of the copper sheet used in the simulation and the fabrication are kept the same to realize the actual antenna design. The reflection coefficient has been measured in the laboratory by using the Agilent vector network analyzer (VNA) N5230. The simulated and measured values of the reflection coefficients of the proposed antenna are plotted in Figure 8.34, and it can be seen from the figure that the simulated values agree well with the experimental values in the operating frequency range.

A small deviation in the impedance bandwidth is observed because of the fabrication imperfection and the irregular surface of the metal sheet. The proposed antenna gives an impedance bandwidth of 60.37% from 3.71 to 6.91 GHz, confirming the wideband characteristics of the MED antenna. Furthermore, the antenna has achieved CP characteristics due to the insertion of I-shaped slots in an extended manner on both sides of the semicircular patches. The axial ratio is the ratio of two orthogonal components of the E-field in a given direction and is often used for the antennas in which the desired polarization is circular. The circularly polarized waves can be generated when two orthogonal E-field components with equal amplitude but with in-phase quadrature are radiated. The acceptable value of the axial ratio is less than 3 dB for the patch antennas [39]. Figure 8.35 presents the variation of the measured and simulated AR values with the frequency. It is seen from the figure that a 3 dB AR bandwidth of 20.60% in the frequency range of 3.71 and 4.55 GHz is achieved in the simulation.

However, there is a small deviation in the measured AR bandwidth because of the inaccuracy in measurement, cutting, and finishing. The gain in the antenna along with radiation patterns were measured in an anechoic chamber by using the standard substitution method. After the analysis of outcomes, it is found that the gain is almost constant over the entire CP band as shown in Figure 8.36.

In general, errors that occur during the measurement is approx. ±0.30 dB, and those further can be minimized by using precision handling processes and precisely calibrated instruments. Simulated and measured results have confirmed that the stable gain of 6 ± 0.5 dBic is accomplished in the operating frequency range of 3.71–6.91 GHz as shown in Figure 8.32. A very small variation has been seen due to errors in anechoic chamber measurement. It is confirmed from Figure 8.37 that the measured radiation efficiency matched well with the simulated result and it is more than 80% in the operating frequency range.

To verify the CP behavior of the proposed antenna, the simulated current distributions using the simulator Ansys HFSS at different time periods are shown in Figure 8.38. Here, T is the time period of the signal at operating frequency.

Due to the symmetrical structure, the surface current distributions of the semicircular patch of only one side has been considered for a clear visibility of the vector currents and the same pattern can be viewed on the other side also. At time t = 0, the surface currents on the defected semi-circular surface flow along the direction of θ = 0°, while the direction of the surface currents has changed in the upward direction at t = T/4 in the direction of θ = 90°. Further, the direction of the vector currents is changed to θ = 180° and θ = 270° at time t = T/2 and 3T/4, respectively. It is clear from the current distribution pattern that the direction of vector currents at t = T/2 and 3T/4 are just opposite to the currents at t = 0 and t = T/4, respectively. It is also confirmed that the surface currents on the electric dipole rotate in the anticlockwise direction to authenticate the right-hand circular polarization (RHCP) behavior of the antenna.

To compare the radiation patterns in the xoz (Φ = 0°) and yoz (Φ = 90°) planes, the radiation patterns obtained from simulation and measurement are plotted for the broadside direction at the center frequencies of the CP band and is shown in Figure 8.39a and b, respectively. The obtained results validate the basic characteristic of the ME dipole antenna, as the radiation patterns from simulation and measurement are unidirectional and symmetric in the E-plane and H-plane. Also, it is clearly visible from Fig 8.39 (b) that the magnitude difference between LHCP and RHCP is more than 3 dB, which is a remarkable point to validate the CP behavior of the antenna and confirming the RHCP waves. The experimental results agree well with the simulated results. Moreover, the radiation patterns of the xoz- plane and the yoz-plane are stable and symmetrical.